1. Field of the Invention
The present invention relates to a controller for a power converter constituted by multiplexed converters, and more particularly, to a controller for a multi-level-output power converter capable of outputting four or more voltage levels, which is able to select an output vector closest to a power command vector at high speed and execute pulse width modulation (hereinafter, called PWM) control by spatial vector comparison at high speed, and which comprises means for implementing switching such that the output voltage is controlled to a sinusoidal wave form.
2. Description of the Related Art
FIG. 1 is a block diagram showing an example of the main circuit composition of a power converter comprising multiplexed (in the diagram, tour) voltage-type converters 5, 6, 7, 8. FIG. 2 gives circuit diagrams showing the composition of each converter 5-8.
In FIG. 1 and FIG. 2, the main circuit of the power converter comprises bridge-connected self-extinguishing type switching elements 9-32, and diodes D9-D32, which are connected respectively in antiparallel to each of the switching elements 9-32. A common DC power supply 4 is connected to the input side of each converter 5-8. 3-phase converters 133-136 are connected respectively to the output side of each converter 5-8, the secondary sides of the 3-phase converters 133-136 being connected mutually and also coupled via a successive resistance 137 and inductance to an AC load 139, such as an AC (motor, or the like.
The following description involves an example where gate turn off thyristors brow are used for the self-extinguishing type switching elements 9-32.
Furthermore, each combination of a GTO and a diode D connected in antiparallel to the GTO is termed an "arm". For example, the arm constituted by GTO9 and 9 is called arm 9, and the arm constituted by GTO10 and D10 is called arm 10. Depending on the direction of the AC current in each converter, current may flow on the diode side, even if the GTO is supplied with an ON gate. Whichever side the current flows, the arm is said to be "switched on".
Furthermore, in addition to the composition illustrated in FIG. 1, wherein DC is taken as a power supply 4 for the power converter consisting of multiplexed voltage-type converters, and the AC side comprises inverters carrying a load, a composition may also be adopted, wherein the AC side is taken as a power supply and the DC side comprises converters connected to a load. Since these compositions differ only with respect to terminology, and share many features relating to operation and technical problems, in the following description, both are treated as multiplex voltage-type converters, without a distinction being made therebetween.
On the other hand, in the prior art, there has been proposed a controller for such a power converter consisting of multiplexed voltage-type converters 5-8, which controls the ignition(firing) state of the self-extinguishing type switching elements in accordance with voltage commands.
FIG. 4 is a block diagram showing the function and sequence of a controller for a power converter of this kind.
Before describing the controller for a power converter illustrated in FIG. 4, an explanation will be given with regard to the voltage vectors which can be generated on the AC side by the four multiplexed voltage-type converters in FIG. 1.
FIG. 3 gives an illustration of output vectors which can be generated by the four multiplexed voltage-type converters.
In FIG. 3, the co-ordinate axes are indicated by U, V, W. For example, it arm 9, arm 10 and arm 11 of converter 5 are switched on, or if arm 12, arm 13 and arm 14 of converter 5 are switched on, the output voltage is zero and the voltage vector in this case is V0.
However, if arms 9, 13 and 14 are switched on, the voltage vector will be V1.
If arms 9, 10 and 14 are switched on, the voltage vector will be V2.
If arms 10, 12 and 14 are switched on, the voltage vector will be V3.
If arms 10, 11 and 12 are switched on, the voltage vector will be V4.
If arms 11, 12 and 13 are switched on, the voltage vector will be V5.
If arms 9, 11 and 13 are switched on, the voltage vector will be V6.
As described above, a single converter is capable of generating seven types of voltage vector: V0, V1, V2, V3, V4, V5 and V6.
FIG. 3 shows all the voltage vectors that can be generated by the four multiplexed voltage-type converters depicted in FIG. 2, and by using a combination of the four unit converters 5-8, it is possible to generate 61 different voltage vectors.
For example, vector V111 represents a stare where three of the converters are generating voltage vector V1 and one of the converters is generating vector V0. In the case of four multiplexed voltage-type converters, this may be written as V0111 or V1110, but for the sake of convenience, the foregoing notation is used.
Correspondingly, the vector 6611 represents a state where two converters are generating vector V6 and two converters are generating vector V1. This applies similarly to the other vectors.
Conventionally, when a voltage command vector for the AC output voltage of a power converter is supplied, an output vector which is closest to the voltage command vector is selected and this is generated by the power converter. In order to achieve this, in the prior art, a controller as illustrated in FIG. 4 has been proposed.
In FIG. 4, symbol 45 denotes an AC voltage command value generator, which generates AC voltage command values, and supplies the voltage command vector for the output voltage of a converter as components RVU1, RVV1 and RVW1 with respect to U, V, W co-ordinates. Here, RV stands for "Reference Voltage".
Symbol 46 denotes a 3-phase .quadrature. 2-phase converter comprising an adder and a multiplier, which converts the components RVU1, RVV1, RVW1 based on U, V, W co-ordinates to components RVA1, RVB1 based on A, B co-ordinates, by means of the calculations: EQU RVA1=RVU1-(RVV1+RVW1)/2 EQU RVB1=(RVV1-RVW1) * 1.732/2
Here, the A axis is parallel to the U axis, and the B axis is advanced by 90.degree. from the A axis.
Furthermore, symbol 168 denotes an outputtable vector generator, which supplies all the output vector values which can be generated by the multiplexed converters, in the form of A axis, B axis co-ordinate value (VnA, VnB). Symbol 170, on the other hand, denotes a vector deviation detection circuit, which calculates the deviation between the ##EQU1## voltage command vector and the output vectors by means of the equation.
Moreover, 171 denotes a comparison and selection circuit, which compares the vector deviations calculated by vector deviation detection circuit 170 and selects the output vector Vn corresponding to the smaller of these.
Symbol 53 denotes a vector.fwdarw.3-phase converter, which generates a GTO switching pattern corresponding to the output vector Vn selected by comparison and selection circuit 171.
Furthermore, symbol 54 denotes a gate signal generator circuit, which generates GTO ignition pulses for multiplex converter 55 constituted by converters 5-8 in FIG. 2.
However, in the control device described above, in the case of four multiplexed converters, for example, only V0, V1, V2, V11, V12, V22, V111, V112, V122, V222, V1111, V1112, V1122, V1222, V2222 in the 0-60.degree. angle range of the spatial vector diagram are taken as output vectors, and in the remaining range from 60-360.degree., it is necessary to calculate the deviation between the 15 output vectors and the voltage command vector, even it the angle is restored to the 0-60.degree. range by rotation through an integral factor of -60.degree..
In this case, the calculation of the vector deviation involves multiplication and square root derivation, and the circuitry becomes extremely large in size if 15 multiplication, square root derivation and comparison circuits are implemented by hardware.
Furthermore, it these functions are implemented in a microcomputer or DSP (digital signal processor, the calculation time will be extremely long.
Moreover, a distinctive feature of spatial vector comparison PWM is that a low distortion factor, low switching loss and high-speed response are achieved by selecting the output vector closest to the voltage command vector at any one instant, but since the selection of the output vector takes time, there is a risk that performance will decline accordingly.
Moreover, if the multiplex number is increased, the number of output vectors will rise in proportion to the square of the multiplex number, thus presenting a problem in that calculation will take even more time.
As described above, in a prior art controller for a power converter, there has been a problem in that output vector selection requires a large amount of time, and hence performance is poor.
Next, a current-type converter is described. FIG. 5 is a diagram of the composition of a multiplex current-type converter.
In FIG. 5, converters A5-A8 for converting DC power to AC power are connected via capacitors A2-A4 to an AC load A1, their respective AC terminals being connected commonly to the AC load A1 such that they operate in parallel. Capacitors A2-A4 are used for absorbing switching surges in the converters A5-A8, and the converters A5-A8 are constituted by self-extinguishing type switching elements A9-A32. The following description involves a case where gate turn off thyristors (GTC) are used as the self-extinguishing type switching elements. DC power supplies A41-A44 are connected to converters A5-A8 via DC reactors A33-A40 which smooth the DC current. Here, it is assumed that DC power supplies A41-A44 are controlled uniformly.
A control circuit as illustrated in FIG. 6 has been proposed as a means for controlling the ignition state of the elements in such a multiplex current-type converter, as disclosed in the prior art in Japanese Laid-Open Patent Hei No. 7-135776.
Firstly, before describing FIG. 6, the current vectors which can be generated on the AC side by the four multiplexed converters in FIG. 5 are described with reference to FIG. 7.
In FIG. 7, the co-ordinate axes are represented by U, V and W. For example, if GTO A9 and GTO A12 of converter A5 are switched on and current is flowing from the U phase to the X phase, then the output current will be zero, and this is taken as I0. The current vector is taken as I1 when GTO A9 and GTO A14 are switched on and current is flowing from the U phase to Z phase; the current vector is taken as I2 when GTO A10 and GTO A14 are switched on and current is flowing from the V phase to Z phase; the current vector is taken as 13 when GTO A10 and GTO A12 are switched on and current is flowing from the V phase to X phase; the current vector is taken as I4 when STO A11 and GTO A12 are switched on and current is flowing from the W phase to X phase; the current vector is taken as is when GTO A11 and GTO A13 are switched on and current is flowing from the W phase to Y phase; and the current vector is taken as I6 when GTO A9 and CTO A13 are switched on and current is flowing from the U phase to Y phase. In this way, a single converter is capable of generating seven current vectors: I0, I1, I2, I3, I4, I5, I6. FIG. 7 shows all the current vectors that can be generated by the four multiplexed current-type converters illustrated in FIG. 5, and by using a combination of four unit converters, it is possible to generate 61 different current vectors. For example, vector I111 represents a state where three of the converters are generating current vector I1 and one converter is generating vector 10. Vector I6611 represents a state where two converters are generating vector I6 and two converters are generating vector I1. This applies similarly to the other vectors.
The essence of a prior art multiplex converter is that when a command vector for the AC output current of the converter is supplied, the outputtable current vector which is closest to the command value vector is selected and this is generated by the converter. As illustrated in FIG. 7, selecting the outputtable vector which is closest to the command value vector is equivalent to splitting the spatial vector diagram into hexagonal domains, each surrounding one of the outputtable vectors, and selecting the vector corresponding to the domain in which the command value vector is located. The control circuit shown in FIG. 6 operates in accordance with a vector selection algorithm based on this domain splitting.
In FIG. 6, the AC current command value generator A45 supplies the output current command value vector of the converter in the form of components RIU1, RIV1 and R1W1 based on U, V, W co-ordinates. The 3-phase.fwdarw.2-phase converter A46 constituted by an adder and multiplier converts the components RIU1, RIV1 and R1W1 based on U, V, W co-ordinates to components RIA1, RIB1 based on A, B co-ordinates, by means of the following calculations. EQU RIA1=RIU1-(RIV1+RIW1)/2 EQU RIB1=(RIV1-RIW1)*1.732/2
Here, the A axis is parallel to the U axis and the B axis is advanced by 90.degree. with respect to the A axis.
Outputtable vector generator A47 supplies all the outputtable vector values which can be generated by the multiplexed converters in the form, of A axis, B axis co-ordinate values (InA,InB). Domain splitting spatial vector diagram generator A48 generates a spatial vector diagram which is split into domains as illustrated in FIG. 7. Domain judgement and vector selection circuit A49 judges which domain of the domain splitting spatial vector diagram contains the command value vector (RIA1,RIB1) and selects the output vector In corresponding to this domain, Vector.fwdarw.3-phase converter A50 generates a GTO switching pattern corresponding to the outputtable vector In selected by the domain judgement and vector selection circuit A49. Gate signal generating circuit A51 generates a GTO ignition pulse for a multiplex converter A52 constituted by converters A5-A8 shown in FIG. 5.
Moreover, in addition to the composition illustrated in FIG. 5 where DC is taken as the power supply and the AC side comprises inverters carrying a load, a composition may also be adopted for the multiplex current-type converter, wherein the AC side is taken as the power supply and the DC side comprises converters connected to a load. These compositions differ only with respect to terminology, and share many features relating to operation and technical problems. Since the problems to be resolved by the present invention are also the same, both are treated as multiplex current-type converters, without a distinction being made therebetween.
The description below refers to FIG. 8, which extracts the 0-60.degree. angle portion of FIG. 7. Here, an "enlarged" view is given to aid visibility, and the generality of the PWM method described below is not reduced in any way thereby.
FIG. 8 relates to a case where a command vector Ir moves from a domain enclosing I111 to a domain enclosing I112.
When the converter output current changes from I111 to I112, the output current will fluctuate with capacitor and motor lag inductance. In a current type inverter, resonance suppression control is usually provided in order to restrict resonance due to capacitor and motor lag inductance. Resonance suppression control seeks to restrict fluctuations in the output current, and therefore it draws back the command vector, which consequently enters the domain of I111. When the converter outputs the current I111 again, the resonance suppression control will act in the opposite direction, causing the current command vector to move in the direction of I112. After repeating this process a number of times, the current command vector stabilizes at I112. This effect is illustrated in FIG. 9.
When "chattering" of this kind occurs, the number of GTO switching operations increases and the power loss associated with switching becomes large.
One method for reducing chattering is to reduce the gain of the resonance suppression control, but if the gain is reduced too far, the original resonance control function will be inadequate and transient response will also deteriorate.
Next, a multi-level-output power converter will be described.
FIG. 10 is a circuit diagram showing one example of the composition of the main circuit of a multi-level-output power converter, to which the present invention applies.
Here, a 5-level-output 3-phase power converter capable of outputting the voltages +2E, +E, 0, -E, -2E is described.
In FIG. 10, U, V and X denote U-phase, V-phase and W-phase single-phase power converters, respectively.
Moreover, Su1-Su8, Sv1-Sv8, Sw1-Sw8 are self-extinguishing elements; Du1-Du8, Dv1-Dv8, Dw1-Dw8 are diodes connected in antiparallel to the self-extinguishing elements Su1-Su8, Sv1-Sv8, Sw1-Sw8; DCu1-Dcu6, Dcv1-Dcv6, Dcw1-Dcw6 are clamping diodes; E1 is a split voltage source between the first terminal and second terminal of a DC voltage source E; E2 is a split voltage source between the second terminal and third terminal of the DC voltage source E, E3 is a split voltage source between the third terminal and fourth terminal of the DC voltage source E; and E4 is a split voltage source between the fourth terminal and fifth terminal of the DC voltage source E.
In a 5-level-output 3--phase power converter constituted as described above, for example, between the self-extinguishing elements Su4 and Su5 of the U-phase power converter in U, a voltage of +E1+E2 (assuming E1=E2=E3=E4=E, then a voltage level of +2E) is output when the self-extinguishing elements Su1-Su4 are on, a voltage of +E2 (assuming same, then a voltage level of E) is output when then self-extinguishing elements su2-Su5 are on, a voltage of 0 is output when the self-extinguishing elements Su3-Su6 are on, a voltage of -E3 (voltage level of -E) is output when self-extinguishing elements Su4-Su7 are on, and a voltage of -E3-E4 (voltage level of -2E) is output when self-extinguishing elements Su5-Su8 are on, thereby enabling five voltage levels, +2E, +E, G, -E, -2E to be output.
In a five-level output inverter, if the self-extinguishing elements Su1-su5 are on simultaneously, for example, then the DC voltage E1 will short between the self-extinguishing elements Su1-Su5 and the clamping diode, and hence an excessive shorting current will flow through the self-extinguishing elements, causing them to break down.
Therefore, in order to prevent this, self-extinguishing elements Su1 and Su5, Su2 and Su6, Su4 and Su8 are controlled such that they operate inversely with respect to each other.
Similar operations are adopted in the V-phase power converter in V and the W-phase power converter in W.
FIG. 11 is a block diagram showing an example of the composition of a prior art controller for controlling a 5-level-output 3-phase power converter as illustrated in FIG. 10.
In FIG. 11, B11-B13 are current detectors for detecting the 3-phase output currents; B14 is a 3-phase load, such as a motor; and B15 is a current controlling circuit, which calculates voltage commands for the power converter for each phase U-W, from current commands and the output currents detected by the current detectors B11-B13.
B16 is a triangular wave generator circuit; and B17 is a comparison circuit, which compares the output from triangular wave generating circuit B16 with the voltage command represented by the output from current control circuit B15, and outputs a command signal switching the self-extinguishing elements constituting the main circuits of the phase power converters U-W on or off. Furthermore, B18 is a gate pulse generator circuit, which uses the command signal from comparison circuit B17 to generate a gate pulse signal for switching the self-extinguishing elements in the phase converters U-W on and off.
FIG. 12 is a wave diagram of a case where a 5-level-output 3-phase power converter is controlled by a control device as shown in FIG. 11.
The following description relates to a case where a 5-level-output 3-phase power converter is controlled by a conventional triangular wave comparison PWM method.
The torque current command corresponding to the 3-phase load B14 in FIG. 11 is taken as the q axis current command Iq*. By performing a 3-phase.fwdarw.2-phase conversion with respect to the output currents Iu, Iv, Iw of phases U, V, W detected by current detectors B11-B13, the final q axis current Iq and d axis current Id are calculated. Current control is applied to the q axis current Tq and the d axis current Id, such that they follow the command values.
Thereby, since the q axis voltage command Vq* and d axis voltage command Vd* are derived, a 2-phase.fwdarw.3-phase conversion is performed to calculate the 3-phase voltage commands Vu*, Vv*, Vw*. A gate signal is obtained by comparing these 3-phase voltage commands Vu*, Vv*, Vw* with the triangular wave carrier signals CAR1-CAR4 from the triangular wave generator 16.
In other words, in the case of the U-phase voltage command Vu*, if Vu*&gt;CAR1, then a gate signal, whereby self-extinguishing elements Su5, Su2, Su3 and Su4 are switched on and self-extinguishing elements Su1, Su6, Su7 and Su8 are switched off, is output.
If CAR1&gt;Vu*&gt;CAR2, then a gate signal, whereby self-extinguishing elements Su2, Su3, Su4 and Su5 are switched on and self-extinguishing elements Su1, Su6, Su7 and Su8 are switched off, is output.
If CAR2&gt;Vu*&gt;CAR3, then a gate signal, whereby self-extinguishing elements Su3, Su4, Su5, Su6 are switched on and self-extinguishing elements Su1, Su2, Su7, Su8 are switched off, is output.
If CAR3&gt;Vu*&gt;CAR4, then a gate signal, whereby self-extinguishing elements Su4. Su5, Su6, 5u7 are switched on and self-extinguishing elements Su1, Su2, Su3, Su8 are switched off, is output.
If CAR4&gt;Vu*, then a gate signal, whereby self-extinguishing elements Su5, Su6, Su7, Su8 are switched on and self-extinguishing elements Su1, Su2, Su3, Su4 are switched off, is output.
Similar switching control to that for the U phase is implemented for the V phase and W phase also.
By implementing switching control as described above, a 3-phase output voltage Vu, Vv, Vw having five levels, +2E, +E, 0, -E, -2E, can be obtained.
FIG. 13 is a circuit diagram showing a further example of the composition of the rain circuit of a multi-level-output 3-phase power converter, to which the present invention applies.
This illustration relates to 7-level-output 3-phase power converter which combines unit cell inverters.
A 7-level-output 3-phase power converter having the composition shown in FIG. 13 is described in P. W. Hammond: "A New Approach to Enhance Power Quality for Medium Voltage AC Drive", IEEE trans. on I.A., 1997.
FIG. 13, B201 is a 3-phase AC power supply; B202 is a power supply switch, such as a contactor, circuit breaker, or the like, for switching AC supply B201 on and off; B203 is a transformer having 9 secondary windings and B204 is a 3-phase multiplexed power converter, which comprises multiplexed single-phase converters 4U, 4V, 4W, each containing three unit cell inverters connected in series, linked by a Y connection
B205 is an AC motor forming a load to which power is supplied by the 3-phase multiplex converter B204.
4U1, 4U2, 4U3, 4V1, 4V2, 4V3, 4W1, 4w2, 4W3 are the unit cell inverters forming the basic constituent elements of 3-phase multiplex converter B204,
FIG. 14 is a circuit diagram showing a detailed compositional example of a unit cell inverter as illustrated in FIG. 13.
In FIG. 14, B301R, B301S, B301T are AC input terminals for the 3-phase power supply; B302 is a standard diode converter comprising six diodes D1-D6; B303 is a filter capacitor for smoothing the DC power; B304 is a single phase inverter comprising four on/off switchable transistors S1-S4; and B305P, B305N are AC output terminals for inverter B304.
In other words, the 3-phase AC input to AC input terminals B301R. B301S, 3301T is converted to DC by the diode converter B302 and this DC is smoothed by filter capacitor B303, thereby yielding a virtually uniform DC voltage E. The output voltage is obtained by acquiring voltage commands by similar means to those for acquiring the 3-phase voltage commands Vu*, Vv*, Vw* in the aforementioned 5-level-output inverter, and carrying out PWM control by triangular wave comparison.
FIG. 15 is a timing chart showing one example of transistor switching and output voltage for a unit cell inverter as illustrated in FIG. 13.
As shown in FIG. 15, by means of the transistors in single-phase inverter B304 comparing CAR1, which is a triangular wave comparison PWM signal, with voltage command V*, on signals are obtained at S1 and S3, and furthermore, by comparing CAR2, which is displaced by a phase of 180.degree. with respect to CAR1, with voltage command V*, on signals are obtained at S2 and S4.
By a combination of transistors switching on, the voltages given in the following table can be obtained between the AC output terminals B305P, B305N.
______________________________________ S1 S3 S2 S4 ______________________________________ +E V* &gt; CAR1 on off V* &gt; CAR2 off on 0 on off CAR2 &gt; V* on off 0 CAR1 &gt; V* off on V* &gt; CAR2 off on -E off on CAR2 &gt; V* on off ______________________________________
By implementing switching control in the aforementioned manner, output voltages of the levels +E, 0 and -E can be obtained for the unit cell inverter illustrated in FIG. 14.
In FIG. 13, a 3-phase multiplex power converter B204 is achieved by means of a composition wherein 3-phase AC power is supplied to each unit cell inverter 4U1-4W3 from six isolated secondary windings in transformer B203, and multiplexed single-phase converters 4U, 4V, 4W, each containing three unit cell inverters 4U1-4W3 of each phase whose output terminals are connected in series, are linked in a Y connection.
Each phase output voltage from the multiplexed single-phase power converters 4U, 4V, 4W forms a voltage comprising the sum of the output voltages from the three unit cell inverters, for example, 4U1-4U3 in the case of the U phase, thereby enabling output voltages of seven levels, namely, +3E, +2E, +E, 0, -E, -2E, -3E, to be obtained.
This description relating to the U phase applies similarly to the V and W phases also.
As described above, in a conventional multi-level-output power converter, since the number of switching operations of the self-extinguishing elements is determined by the triangular wave carrier frequency, the loss associated with switching is large and the efficiency of the power converter falls.
Furthermore, natural harmonics of the triangular wave carrier frequency are superimposed on the output current of the power converter.